Application Note 50
Implementing the RC5050 and RC5051 DC-DC
®
Converters on Pentium Pro Motherboards
Introduction
Intel Pentium Pro Processor Power
Requirements
This document describes how to implement a switching volt-
age regulator using an RC5050 or an RC5051 high speed
controller, a power inductor, a Schottky diode, appropriate
capacitors, and external power MOSFETs. This regulator
forms a step down DC-DC converter that can deliver up to
14.5A of continuous load current at voltages ranging from
1.3V to 3.5V. A specific application circuit, design consider-
ations, component selection, PCB layout guidelines, and per-
formance evaluations are covered in detail.
®
Refer to Intel’s AP-523 Application Note, Pentium Pro
Processor Power Distribution Guidelines, November 1995
(order number 242764-001), as a basic reference. The speci-
fications contained in this document have been modified
slightly from the original Intel document to include updated
specifications for more recent processors. Please contact
Intel Corporation for specific details.
Input Voltages
In the past 10 years, microprocessors have evolved at such an
exponential rate that a modern chip can rival the computing
power of a mainframe computer. Such evolution has been
possible because of the increasing numbers of transistors that
processors integrate. Pentium CPUs, for example, integrate
well over 5 million transistors on a single piece of silicon.
Available inputs are +12V ±5% and +5V ±5%. Either one or
both of these inputs can be used by the DC-DC converter.
The input voltage requirements for Raytheon’s RC5050
and RC5051 DC-DC converters are listed in Table 1.
Table 1. Input Voltage Requirements
To integrate so many transistors on a piece of silicon, their
physical geometry has been reduced to the sub-micron level.
As a result of each geometry reduction, the corresponding
operational voltage for each transistor has also been reduced.
The changing CPU voltage demands the design of a pro-
grammable power supply—a design that is not completely
re-engineered with every change in CPU voltage.
MOSFET
Drain
MOSFET
Gate Bias
Part #
Vcc for IC
RC5050
RC5051
+5V ±5%
+5V ±5%
12V ±5% or
+5V ±5%
Pentium Pro DC Power Requirements
®
Refer to Table 2, Intel Pentium Pro and OverDrive Proces-
sor Power Specifications. For a motherboard designs without
a standard VRM (Voltage Regulator Module) socket, the
on-board DC-DC converter must supply a minimum of
13.9A of current @2.5V and 12.4A of current @3.3V. For a
Flexible Motherboard design, the on-board DC-DC con-
The voltage range of the CPU has shown a downwards trend
for the past 5 years: from 3.3V for the Pentium, to 3.1V for
the Pentium Pro, and to 1.8V for future processors. With this
trend in mind, Raytheon Electronics has designed the
RC5050 and RC5051 controllers. These controllers integrate
the necessary programmability to address the changing
power supply requirements of lower voltage CPUs.
verter must supply 14.5A maximum I P.
CC
DC Voltage Regulation
Previous generations of DC-DC converter controllers were
designed with fixed output voltages adjustable only with a
set of external resistors. In a high volume production envi-
ronment (such as with personal computers), however, a CPU
voltage change requires a CPU board re-design to accommo-
date the new voltage requirement. The 5-bit DAC in the
RC5050 and the RC5051 reads the voltage ID code that is
programmed into modern processors and provides the appro-
priate CPU voltage. In this manner, the PC board does not
have to be re-designed each time the CPU voltage changes.
The CPU can thus automatically configure its own required
supply voltage.
As indicated in Table 2, the voltage level supplied to the
CPU must be within ±5% of its nominal setting. Voltage reg-
ulation limits must include:
• Output load ranges specified in Table 2
• Output ripple/noise
• DC output initial voltage set point
• Temperature and warm up drift (Ambient +10°C to +50°C
at full load with a maximum rate of change of 5°C per 10
minutes minimum but no more than 10°C per hour)
• Output load transient with:
Slew rate >30A/µs at converter pins
Range: 0.3A - I P Max (as defined in Table 2).
CC
Rev. 1.1.0
APPLICATION NOTE
AN50
I/O Controls
The RC5050 and RC5051 Controllers
In addition to the Voltage Identification, there are several sig-
nals that control the DC-DC converter or provide feedback
from the DC-DC converter to the CPU. They are Power-
Good (PWRGD), Output Enable (OUTEN), and Upgrade
Present (UP#). These signals will be discussed later.
The RC5050 is a programmable non-synchronous DC-DC
controller IC. The RC5051 is a synchronous version of the
RC5050. When designed around the appropriate external
components, either of these devices can be configured to
deliver more than 14.5A of output current. The RC5050 and
RC5051 utilize both current-mode and voltage-mode PWM
control to create an integrated step-down voltage regulator.
The key differences between the RC5050 and RC5051 are
listed in Table 4.
RC5050 and RC5051 Description
Simple Step-Down Converter
Table 4. RC5050 and RC5051 Differences
S1
L1
+
RL Vout
–
RC5051
RC5050
Operation
Package
Synchronous Non-Synchronous
VIN
D1
C1
20-SOIC
Yes
20-SOIC
Yes
Output Enable/
Disable
65-5050-06
Figure 1. Simple Buck DC-DC Converter
Main Control Loop
Refer to the RC5051 Block Diagram illustrated in Figure 2.
The control loop of the regulator contains two main sections;
the analog control block and the digital control block. The
analog section consists of signal conditioning amplifiers
feeding into a set of comparators which provide the inputs to
the digital control block. The signal conditioning section
accepts inputs from the IFB (current feedback) and VFB
(voltage feedback) pins and sets up two controlling signal
paths. The voltage control path amplifies the VFB signal and
presents the output to one of the summing amplifier inputs.
The current control path takes the difference between the
IFB and VFB pins and presents the resulting signal to
another input of the summing amplifier. These two signals
are then summed together with the slope compensation input
from the oscillator. This output is then presented to a
comparator, which provides the main PWM control signal to
the digital control block.
Figure 1 illustrates a step-down DC-DC converter with no
feedback control. The derivation of the basic step-down con-
verter is the basis for the design equations for the RC5050
and RC5051. Referring to Figure 1, the basic operation
begins by closing the switch S1. When S1 is closed, the input
voltage V is impressed across inductor L1. The current
IN
flowing in this inductor is given by the following equation:
(VIN – VOUT)TON
IL = ----------------------------------------------
L1
where T is the duty cycle (the time when S1 is closed).
ON
When S1 opens, the diode D1 conducts the inductor
current and the output current is delivered to the load accord-
ing to the following equation:
VOUT(TS – TON
IL = -------------------------------------------
L1
)
The additional comparators in the analog control section set
the point at which the current limit comparator disables the
output drive signals to the external power MOSFETs.
whereT is the overall switching period and (T - T ) is the
time during which S1 is open.
S
S
ON
The digital control block takes the comparator inputs and the
main clock signal from the oscillator to provide the appropri-
ate pulses to the HIDRV and LODRV output pins. These
pins control the external power MOSFETs. The digital sec-
tion utilizes high speed Schottky transistor logic, allowing
the RC5050 and the RC5051 to operate at clock speeds as
high as 1MHz.
By solving these two equations, we can arrive at the basic
relationship for the output voltage of a step-down converter:
TON
----------
VOUT = VIN
TS
In order to obtain a more accurate approximation for V
,
OUT
we must also include the forward voltage V across diode
D
D1 and the switching loss, V . After taking into account
these factors, the new relationship becomes:
SW
High Current Output Drivers
The RC5051 contains two identical high current output
drivers that utilize high speed bipolar transistors in a
push-pull configuration. Each driver is capable of
delivering 1A of current in less than 100ns. Each driver’s
power and ground are separated from the chip’s power and
ground for additional switching noise immunity.
TON
----------
– VD
VOUT = (VIN + VD – VSW
)
TS
where V
= MOSFET switching loss
SW
= I • R
L
DS,ON
3
AN50
APPLICATION NOTE
+12V
RC5051
+5V
OSC
–
+
–
+
–
+
DIGITAL
CONTROL
VO
–
+
1.24v
REFERENCE
5-BIT
DAC
VREF
POWER
GOOD
PWRGD
65-5051-01
VID0 VID2 RSEL
VID1 VID3
Figure 2. RC5051 Block Diagram
The HIDRV driver has a power supply, VCCQP, supplied
from a 12V source as illustrated in Figure 2. The resulting
voltage is sufficient to provide the gate to source voltage to
the external MOSFET that is required to achieve a low
age and outputs an active-low interrupt signal to the CPU
when the power supply voltage exceeds ±12% of nominal.
The Power Good flag provides no other control function to
the RC5050 or the RC5051.
R
. Since the low side synchronous FET is referenced
DS,ON
to ground, there is no need to boost the gate drive voltage,
and its VCCP power pin can be tied to VCC.
Output Enable (OUTEN)
The DC-DC converter accepts an open collector signal for
controlling the output voltage. The low state disables the out-
put voltage. When disabled, the PWRGD output is in the low
state.
Internal Voltage Reference
The reference included in the RC5050 and RC5051 is a pre-
cision band-gap voltage reference. The internal resistors are
precisely trimmed to provide a near zero temperature coeffi-
cient (TC). Added to the reference input is the resulting out-
put from an integrated 5-bit DAC—provided in accordance
to the Pentium Pro specification guidelines. These guidelines
require the DC-DC converter output to be directly program-
mable via a 4-bit voltage identification (VID) code. This
code scales the reference voltage from 2.0V (no CPU) to
3.5V in 100mV increments. To target future generations of
low-voltage processors, the RC5050 and RC5051 incorpo-
rate a VID4 pin to allow additional programmability between
1.3V and 2.05V. For guaranteed stable operation under all
operating conditions, a 0.1µF of decoupling capacitance
should be connected to the VREF pin. No load should be
imposed on this pin.
Upgrade Present (UP#)
Intel specifications state that the DC-DC converter should
accept an open collector signal, used to indicate the presence
of an upgrade processor. The typical state is high (that is, a
standard processor is in the system). When in the low or
ground state (an OverDrive processor is present), the output
voltage must be disabled unless the converter can supply the
requirements of the OverDrive processor. When disabled, the
PWRGD output must be in the low state. Because the
RC5050 and RC5051 can supply the requirements of the
OverDrive processor, the #UP signal is not required.
Over-Voltage Protection
The RC5050 and RC5051 constantly monitor the output
voltage for protection against over voltage conditions. If the
voltage at the VFB pin exceeds 20% of the selected program
voltage, an over-voltage condition is assumed and the chip
disables the output drive signal to the external MOSFET(s).
Power Good (PWRGD)
The RC5050 and RC5051 Power Good function is designed
in accordance with the Pentium Pro DC-DC converter speci-
fication to provide a constant voltage monitor on the VFB
pin. The circuit compares the VFB signal to the VREF volt-
4
APPLICATION NOTE
AN50
In general, a lower operating frequency decreases the peak
Short Circuit Protection
ripple current flowing in the output inductor, thus allowing
the use of a smaller inductor value. Unfortunately, operation
at lower frequencies increases the amount of energy storage
that must be provided by the bulk output capacitors during
load transients due to slower loop response of the controller.
A current sense methodology is implemented to disable the
output drive signal to the MOSFET(s) when an over-current
condition is detected. The voltage drop created by the output
current flowing across a sense resistor is presented to an
internal comparator. When the voltage developed across the
sense resistor exceeds the comparator threshold voltage, the
chip reduces the output drive signal to the MOSFET(s).
In addition, the efficiency losses due to switching of the
MOSFETs increase as the operating frequency is increased.
Thus, efficiency is optimized at lower operating frequencies.
An operating frequency of 300 kHz was chosen to optimize
efficiency while maintaining excellent regulation and tran-
sient performance under all operating conditions.
The DC-DC converter returns to normal operation after the
fault has been removed, for either an over-voltage or a short
circuit condition.
Oscillator
Design Considerations and
Component Selection
Figure 3 shows a typical non-synchronous application using
the RC5050. Figure 4 illustrates the synchronous applica-
tion using the RC5051.
The RC5050 and RC5051 oscillator section uses a fixed cur-
rent capacitor charging configuration. An external capacitor
(C
) is used to preset the oscillator frequency between
EXT
200KHz and 1MHz. This scheme allows maximum flexibil-
ity in setting the switching frequency and in choosing exter-
nal components.
+12V
L2
+5V
2.5µH
C5
C4
C3
C1
C2
R5
47
0.1µF
0.1µF
1000 µF 1000 µF
1000 µF
D1
C9
C8
1N4691
0.1µF
0.1µF
M1
M2
C12
IRF7413
11
12
13
14
15
10
1µF
9
8
7
6
5
IRF7413
RSENSE
L1
VO
1.3µ H
6mΩ
C6
RC5050
4.7µF
16
17
18
19
20
VREF
GND
DS1
4
3
2
1
MBR2015CTL
C7
0.1µF
CEXT
100pF
VID4
VID3
VCC
R6
10K
VID2
VID1
VID0
PWRGD
C11
0.1µF
ENABLE
C10
0.1µF
Figure 3. Non-Synchronous DC-DC Converter Application Schematic Using the RC5050
5
AN50
APPLICATION NOTE
+12V
+5V
L2
2.5µH
C5
C4
C3
C1
C2
R5
47
0.1µF
0.1µF
1000 µF 1000 µF
1000 µF
D1
C9
C8
1N4691
0.1µF
0.1µF
M1
M2
C12
IRF7413
11
12
13
14
15
10
1µF
9
8
7
6
5
RSENSE
IRF7413
L1
C6
4.7
VO
1.3µ H
6mΩ
F
µ
RC5051
16
17
18
19
20
VREF
DS1
1N5817
4
3
2
1
C7
M3
M4
IRF7413
0.1µF
IRF7413
GND
CEXT
100pF
VID4
VID3
VCC
R6
10K
VID2
VID1
VID0
PWRGD
C11
0.1µF
ENABLE
C10
0.1µF
Figure 4. Synchronous DC-DC Converter Application Schematic Using the RC5051
6
APPLICATION NOTE
AN50
• Power package with low Thermal Resistance
• Drain current rating of 20A minimum
• Drain-Source voltage > 15V.
MOSFET Selection Cosiderations
MOSFET Selection
This application requires N-channel Logic Level Enhance-
ment Mode Field Effect Transistors. Desired characteristics
are as follows:
The on-resistance (R
) is the primary parameter for
DS,ON
MOSFET selection. It determines the power dissipation
within the MOSFET and, therefore, significantly affects the
efficiency of the DC-DC converter. Table 5 is a selection
table for MOSFETs.
• Low Static Drain-Source On-Resistance,
R
< 37 mΩ (lower is better)
DS,ON
• Low gate drive voltage, V ≤ 4.5V
GS
Table 3. MOSFET Selection Table
Manufacturer & Model #
R
(mΩ)
DS,ON
Thermal
P ackage Resistance
1
Conditions
Typ.
25
Max.
Fuji
V
V
V
= 4V, I = 17.5A T = 25°C
37
—
20
34
15
TO-220
Φ
Φ
= 75
= 50
GS
GS
GS
D
J
JA
JA
2SK1388
T = 125°C
37
J
Siliconix
SI4410DY
= 4.5V, I = 5A T = 25°C
16.5
28
SO-8
(SMD)
D
J
T = 125°C
J
National Semiconductor
NDP706AL
= 5V, I = 40A T = 25°C
13
TO-220
Φ
Φ
= 62.5
= 1.5
D
J
JA
JC
NDP706AEL
National Semiconductor
NDP603AL
T = 125°C
20
31
42
22
33
6
24
40
54
25
40
9
J
V
V
V
V
V
V
= 4.5V, I = 10A T = 25°C
TO-220
TO-220
TO-263
Φ
Φ
Φ
Φ
Φ
Φ
Φ
Φ
Φ
Φ
Φ
= 62.5
= 2.5
GS
GS
GS
GS
GS
GS
D
J
JA
JC
JA
JC
JA
JC
JA
JC
JA
JC
JA
T = 125°C
J
National Semiconductor
NDP606AL
= 5V, I = 24A T = 25°C
= 62.5
= 1.5
D
J
T = 125°C
J
Motorola
= 5V, I = 37.5A T = 25°C
= 62.5
= 1.0
D
J
2
MTB75N03HDL
Int. Rectifier
IRLZ44
T = 125°C
9.3
—
—
—
—
14
28
46
19
31
18
(D PAK)
J
= 5V, I = 31A T = 25°C
TO-220
TO-220
= 62.5
= 1.0
D
J
T = 125°C
J
Int. Rectifier
IRL3103S
= 4.5V, I = 28A T = 25°C
= 62.5
= 1.0
D
J
T = 125°C
J
Intl Rectifier
IRF7413
= 4.5V,
T = 25°C
SO-8
SMD
= 50
A
I = 3.7A
D
Note:
1. R
values at Tj = 125°C for most devices were extrapolated from the typical operating curves supplied by the
DS,ON
manufacturers and are approximations only.
7
AN50
APPLICATION NOTE
Two MOSFETs in parallel.
+5V
We recommend two MOSFETs used in parallel instead of
one single MOSFET. The following significant advantages
are realized using two MOSFETs in parallel:
DS2
VCCQP
HIDRV
M1
• Significant reduction of Power dissipation.
Maximum current of 14A with one MOSFET:
L1
RS
CP
VO
PWM/PFM
Control
2
P
= (I R
)(Duty Cycle) =
MOSFET
2
DS,ON
CB
DS1
(14) (0.050*)(3.3+0.4)/(5+0.4-0.35) = 7.2 W
With two MOSFETs in parallel:
65-AP50-01
2
P
= (I R
)(Duty Cycle) =
MOSFET
2
DS,ON
Figure 5. Charge Pump Configuration
• Method 2. 12V Gate Bias.
(14/2) (0.037*)(3.3+0.4)/(5+0.4-0.35) = 1.3W/FET
* Note: R
increases with temperature. Assume R
= 25mΩ at
DS,ON
DS,ON
DS,ON
Figure 6 illustrates how a 12V source can be used to bias
the VCCQP. A 47 Ω resistor is used to limit the transient
current into the VCCQP pin and a 1µF capacitor filter is
used to filter the VCCQP supply. This method provides a
25°C. R
can easily increase to 50mΩ at high temperature when
using a single MOSFET. When using two MOSFETs in parallel, the
temperature effects should not cause the R
listed maximum value of 37mΩ.
to rise above the
DS,ON
• Less heat sink required.
higher gate bias voltage (V ) to the MOSFET, and there-
GS
With power dissipation down to around one watt and with
MOSFETs mounted flat on the motherboard, considerable
less heat sink is required. The junction-to-case thermal
resistance for the MOSFET package (TO-220) is typically
at 2°C/W and the motherboard serves as an excellent heat
sink.
fore reduces the R
of the MOSFET and reduces the
DS,ON
power loss due to the MOSFET. Figure 7 shows how
reduces dramatically with V increases. A 6.2V
R
DS,ON
GS
Zener diode (D1) is placed to clamp the voltage at VCCQP
to a maximum of 12V and ensure that the absolute maxi-
mum voltage of the IC will not be exceeded
• Higher current capability.
+5V
With thermal management under control, this on-board
DC-DC converter is able to deliver load currents up to
14.5A with no performance or reliability concerns.
47Ω
+12V
D1
6.2V
MOSFET Gate Bias
VCCQP
M1
MOSFET can be biased by one of two methods: Charge
Pump and 12V Gate Bias.
HIDRV
L1
RS
1µF
VO
PWM/PFM
Control
• Method 1. Charge pump (or Boostrap) method.
Figure 5 employs a charge pump to provide gate bias.
Capacitor CP is the charge pump deployed to boost the
voltage of the RC5050 output driver. When the MOSFET
switches off, the source of the MOSFET is at -0.6V.
VCCQP is charged through the Schottky diode to 4.5V.
Thus, the capacitor CP is charged to 5V. When the MOS-
FET turns on, the source of the MOSFET voltage is equal
to 5V. The capacitor voltage follows, and hence provides a
voltage at VCCQP equal to 10V. The Schottky diode is
required to provide the charge path when the MOSFET is
off, and reverses bias when the VCCQP goes to 10V. The
charge pump capacitor, CP, needs to be a high Q, high fre-
quency capacitor. A 1µF ceramic capacitor capacitor is
recommended here.
CB
DS1
65-AP50-02
Figure 6. 12V Gate Bias Configuration
0.1
0.09
0.08
0.07
0.06
0.05
0.04
0.03
0.02
0.01
0
R(DS)Fuji
R(DS)7060
R(DS)706A
R(DS)-706AEL
1.5 2 2.5 3 3.5 4
5
6
7
8
9
10 11
Gate-Source Voltage, V (V)
GS
Figure 7. R
vs. V for Selected MOSFETs
GS
DS,ON
8
APPLICATION NOTE
AN50
Converter Efficiency
Losses due to parasitic resistance in the switches, coil, and
sense resistor dominate at high load-current level. The major
loss mechanisms under heavy loads, in usual order of impor-
tance, are:
• gate-charge losses
• diode-conduction losses
• transition losses
• Input Capacitor losses
• losses due to the operating supply current of the IC.
2
• MOSFET I R Losses
• Coil Losses
• Sense Resistor Losses
Calculation of Converter Efficiency Under Heavy Loads
POUT IOUT × VOUT
Efficiency = ------------- = -------------------------------------------------------
pIN IOUT × VOUT + PLOSS
PLOSS = PDMOSFET + PDCOIL + PDSENSER + PDGATE + PDDIODE + PDTRAN + PDCAP + PDIC
V
OUT + VD
where PDMOSFET = IOUT2 × RDS,ON × DutyCycle , where DutyCycle = -----------------------------------------
V
IN + VD – VSW
PDCOIL = IOUT2 × RCOIL
PDSENSER = IOUT2 × RSENSE
PDGATE = qGATE × f × 5V , where q
is the gate charge and f is the switching frequency
GATE
PDDIODE = Vf × ID(1 – Dutycycle)
VIN2 × CRSS × ILOAD × f
PDTRAN = ------------------------------------------------------------- , where C
IDRIVE
is the reverse transfer capacitance of the high-side MOSFET.
RSS
PDCAP = IRMS2 × ESR
PDIC = VCC × ICC
Example
3.3 + 0.5
DutyCycle = ----------------------------- = 0.73
5 + 0.5 – 0.3
PDMOSFET = 102 × 0.030 × 0.73 = 2.19W
PDCOIL = 102 × 0.010 = 1W
PDSENSER = 102 × 0.0065 = 0.65W
PDGATE = CV × f × 5V = 1.75nf × (9 – 1)V × 285Khz × 5V = 0.019W
PDDIODE = 0.5 × 10(1 – 0.73) = 1.35W
52 × 400pf × 10 × 285khz
----------------------------------------------------------------
∼ 0.010W
PDTRAN
=
0.7A
PDCAP = (7.5 – 2.5)2 × 0.015 = 0.37W
PDIC = 0.2W
PDLOSS = 2.19W + 1.0W + 0.65W + 0.019W + 1.35W + 0.010W + 0.37W + 0.2W = 5.789W
3.3 × 10
---------------------------------------
∴ Efficiency =
≈ 85%
3.3 × 10 + 5.815
9
AN50
APPLICATION NOTE
When designing the external current sense circuitry, pay
careful attention to the output limitations during normal
operation and during a fault condition. If the short circuit
protection threshold current is set too low, the DC-DC con-
verter may not be able to continuously deliver the maximum
CPU load current. If the threshold level is too high, the out-
put driver may not be disabled at a safe limit and the result-
ing power dissipation within the MOSFET(s) may rise to
destructive levels.
Selecting the Inductor
The inductor is one of the most critical components to be
selected for a DC-DC converter application. The critical
parameters are inductance (L), maximum DC current (I ),
O
and DC coil resistance (R ). The inductor core material is a
l
crucial factor in determining the amount of current the
inductor is able to withstand. As with all engineering
designs, tradeoffs exist between various types of core materi-
als. In general, Ferrites are popular due to low cost, low EMI
properties, and high frequency (>500KHz) characteristics.
Molypermalloy powder (MPP) materials exhibit good satu-
ration characteristics, low EMI, and low hysteresis losses,
but tend to be expensive and more effectively utilized at
operating frequencies below 400KHz. Another critical
parameter is the DC winding resistance of the inductor. This
value should typically be reduced as much as possible, as the
The following is the design equation used to set the short cir-
cuit threshold limit:
Vth
-------
, where: ISC = Output short circuit current
RSENSE
=
ISC
(Ipk – Imin
)
ISC ≥ Iinductor = ILoad, max + ---------------------------
power loss in the DC resistance degrades the efficiency of
the converter by the relationship: P = I x R . The value
of the inductor is a function of the oscillator duty cycle
2
2
loss
O
l
Where I and I are peak ripple current and
pk
min
I
= maximum output load current.
load, max
(T ) and the maximum inductor current (I ). I can be
ON
PK PK
calculated from the relationship:
You must also take into account the current (I -I ), or the
pk min
ripple current flowing through the inductor under normal
operation. Figure 8 illustrates the inductor current waveform
for the RC5050 DC-DC converter at maximum load.
VIN – VSW – VD
-----------------------------------------
TON
IPK = IMIN
+
L
Where T is the maximum duty cycle and V is the
forward voltage of diode DS1.
ON
D
Ipk
I
(I -I )/2
pk min
Then the inductor value can be calculated using the
relationship:
ILOAD, MAX
t
Imin
VIN – VSW – VO
-----------------------------------------
L =
TON
I
PK – IMIN
TON
TOFF
T=1/f s
Where V (R
x I ) is the drain-to-source voltage of
SW
DS,ON
O
M1 when it is switched on.
Figure 8. Typical DC-DC Converter
Inductor Current Waveform
Implementing Short Circuit Protection
The calculation of this ripple current is as follows:
Intel currently requires all power supply manufacturers to
provide continuous protection against short circuit condi-
tions that may damage the CPU. To address this requirement,
Raytheon Electronics has implemented a current sense meth-
odology to limit the power delivered to the load in the event
of overcurrent. The voltage drop created by the output cur-
rent across a sense resistor is presented to one terminal of an
internal comparator with hysterisis. The other comparator
terminal has the threshold voltage, nominally of 120mV.
Table 6 states the limits for the comparator threshold of the
Switching Regulator.
(VIN – VSW – VOUT
)
(VOUT + VD)
(VIN – VSW + VD)
(Ipk – Imin
)
----------------------------------------------------- ----------------------------------------------
--------------------------- =
2
×
T
L
where:
V
V
= input voltage to converter,
= voltage across switcher MOSFET = I
V = Forward Voltage of the Schottky diode,
T = the switching period of the converter = 1/f , and
IN
x R
,
SW
LOAD
DS,ON
D
S
f = switching frequency.
S
For an input voltage of 5V, output voltage of 3.3V, L equals
Table 6. RC5050 Short Circuit Comparator
Threshold Voltage
1.3µH and a switching frequency of 285KHz (using
C
= 100pF), the inductor current can be calculated at
EXT
approximately 1A:
Short Circuit Comparator
(Ipk – Imin
)
V
(mV)
(5.0 – 14.5 × 0.037 – 3.3)
threshold
-------------------------------------------------------------
--------------------------- =
×
1.3 × 10–6
2
Typical
120
(3.3 + 0.5)
1
Minimum
Maximum
100
--------------------------------------------------------- -----------------------
×
= 2A
285 × 103
5.0 – 14.5 × 0.037 + 0.5
140
10
APPLICATION NOTE
AN50
Therefore, for load current of 14.5A, the peak current
The next step is to determine the value of the sense resistor.
through the inductor, I , is found to be approximately
15.5A:
Including sense resistor tolerance, the sense resistor value
can be approximated as follows
pk
Vth,min
Vth,min
(IPK – Imin
)
----------------
ISC
----------------------------------
1.0 + ILoad,max
RSENSE
=
× (1 – TF) =
× (1 – TF)
ISC ≥ Iinductor = ILoad, max + ----------------------------- = 14.5 + 2 = 16.5A
2
Where TF = Tolerance Factor for the sense resistor.
Therefore, the short circuit detection threshold must be at
least 16.5A.
Table 7 describes tolerance, size, power capability, tempera-
ture coefficient and cost of various type of sense resistors.
Table 7. Comparison of Sense Resistors
Discrete Iron
Discrete Metal
Strip Surface
Mount Resistor
(Dale)
Discrete
CuNi Alloy
Wire Resistor
(Copel)
Discrete MnCu
Alloy Wire
Resistor
Motherboard
Alloy
Description
Trace Resistor
Resistor (IRC)
Tolerance
Factor (TF)
±29%
±5%
(±1% available)
±1%
±10%
±10%
Size
(L x W x H)
2" x 0.2" x 0.001" 0.45" x 0.065" x 0.25" x 0.125" x 0.200" x 0.04" x
0.200" x 0.04" x
0.100"
(1 oz Cu trace)
0.200"
0.025"
0.160"
Power capability
>50A/in
1 watt
1 watt
1 watt
1 watt
(3W and 5W
available)
Temperature
Coefficient
+4,000 ppm
+30 ppm
±75 ppm
±30 ppm
±20 ppm
Cost
@10,000 piece
Low included in
motherboard
$0.31
$0.47
$0.09
$0.09
Refer to Appendix A for Directory of component suppliers
Based on the Tolerance Factor in the above table, for an
Table 8. R
for Various Load Currents
sense
embedded PC trace resistor and for I
= 14.5A:
load,max
R
R
SENSE
SENSE
Vth,min
I
PC Trace
Discrete
Load,max
----------------------------------------
× (1 – TF) =
RSENSE
=
2.0A + ILoad, max
(A)
Resistor (mΩ)
Resistor (mΩ)
10.0
11.2
12.4
13.9
14.0
14.5
5.9
5.4
4.9
4.5
4.4
4.3
7.9
7.2
6.6
6.0
5.9
5.8
100mV
---------------------------------
× (1 – 29%) = 4.3mΩ
2.0A + 14.5A
For a discrete resistor and for I
Vth,min
= 14.5A:
load, max
----------------------------------------
× (1 – TF) =
RSENSE
=
2.0A + ILoad, max
100mV
---------------------------------
× (1 – 5%) = 5.8mΩ
Discrete Sense Resistor
2.0A + 14.5A
Discrete Iron Alloy resistors come in variety of tolerances
and power ratings, and are most ideal for precision imple-
mentation. MnCu Alloy wire resistors or CuNi Alloy wire
resistors are ideal for low cost implementations.
For user convenience, Table 8 lists the recommended values
for sense resistors for various load currents using embedded
PC trace resistors and discrete resistors.
11
AN50
APPLICATION NOTE
Embedded Sense Resistor (PC Trace Resistor)
where:
Embedded PC trace resistors have the advantage of near zero
cost implementation. However, the value of the PC trace
resistor has large variations. Embedded resistors have 3
major error sources: the sheet resistivity of the inner layer,
the mismatch due to L/W, and the temperature variation of
the resistor. All three error sources must be considered for
laying out embedded sense resistors.
ρ = Resistivity(µΩ-mil),
L = Length(mils),
W = Width(mils), and
t = Thickness(mils).
L
W
t
For 1oz copper, t = 1.35 mils, ρ = 717.86 µΩ-mil,
1 L/1 W = 1 Square ( ■ ).
For example, you can layout a 5.30mΩ embedded sense
• Sheet resistivity.
resistor using the equations above:
For 1 ounce copper, the thickness variation is typically
1.15 mil to 1.35 mil. Therefore error due to sheet resistiv-
ity is (1.35 - 1.15)/1.25 = 16%
IL
10
W = --------- = --------- = 200mils
0.05
0.05
R × W × t
0.00530 × 200 × 1.35
L = ----------------------- = --------------------------------------------------- = 2000mils
• Mismatch due to L/W.
ρ
717.86
Percent error in L/W is dictated by geometry and the
power dissipation capability of the sense resistor. The
sense resistor must be able to handle the load current and
therefore requires a minimum width which is calculated as
follows.
L/W = 10 ■
Therefore, to model 5.30mΩ embedded sense resistor, you
need W = 200 mils and L = 2000 mils. Refer to Figure 9.
IL
W = ---------
0.05
1
1
1
1
1
1
1
1
1
1
W = 200 mils
where: W = minimum width required for proper power
dissipation (mils) and I = Load Current in Amps.
L = 2000
Figure 9. 5.30mΩ Sense Resistor (10 ■)
L
For 15A of load current, minimum width required is
300mils, which reflects a 1% L/W error.
You can also implement the sense resistor in the following
manner. Each corner square is counted as 0.6 square since
current flowing through the corner square does not flow
uniformly and it is concentrated towards the inside edge, as
shown in Figure 10.
• Thermal Consideration.
2
Due to I R power losses the surface temperature of the
resistor will increase leading to a higher value. In addition,
ambient temperature
variation will add the change in resistor value:
1
1
1
1
1
1
R = R20[1 + α20(T – 20)]
.6
.6
1
1
where: R is the resistance at 20°C, α20 = 0.00393/ °C, T
20
.8
is the operating temperature, and R is the desired value.
Figure 10. 5.30mΩ Sense Resistor (10 ■)
For temperature T = 50°C, the %R change = 12%.
A Design Example Combining an Embedded Resistor
and a Discrete Resistor
Table 9 is the summary of the tolerance for the Embedded
PC Trace Resistor.
For low cost implementation, the embedded PC trace resistor
is most desirable. However, its wide tolerance (29%) pre-
sents a challenge. In addition, requirements for the CPU
change frequently, and, thus, the maximum load current may
be subject to change. Combining embedded resistors with
discrete resistors may be a desirable option. Figure 11 shows
a design that provides flexibility with a solution to address
wide tolerances.
Table 9. Summary PC Trace Resistor Tolerance
Tolerance due to Sheet Resistivity variation
Tolerance due to L/W error
16%
1%
Tolerance due to temperature variation
Total Tolerance for PC Trace Resistor
12%
29%
In this design, you have the option to choose an embedded
or a discrete MnCu sense resistor. To use the discrete sense
resistor, populate R21 with a shorting bar (zero Ohm resis-
tor) for proper Kelvin connection and add the MnCu sense
resistor. To use the embedded sense resistor, on the other
hand, populate R22 with a shorting bar for Kelvin connec-
Design Rules for Using an Embedded Resistor
The basic equation for laying an embedded resistor is:
L
W × t
-------------
R = ρ ×
12
APPLICATION NOTE
AN50
Embedded Sense Resistor
IFBH
MnCu Discrete
Resistor
R21
R22
IFBL
Output Power
Plane (Vout)
R-∆r
R
R+∆r
Figure 11. Short Circuit Sense Resistor Design Using a PC Trace Resistor and an Optional Discrete Sense Resistor
tion. The embedded sense resistor allows the user to choose a
plus or a minus delta resistance tap to offset any large sheet
resistivity change. In this design, the center tap yields 6mΩ,
the left tap yields 6.7mΩ, and the right tap yields 5.3mΩ.
Power Dissipation Consideration During a
Short Circuit Condition
The RC5050 and RC5051 controllers respond to an output
short circuit by drastically changing the duty cycle of the
gate drive signal to the power MOSFET. In doing this, the
power MOSFET is protected from stress and from eventual
failure. Figure 13A shows the gate drive signal of a typical
RC5050 operating in continuous mode with a load current of
10A. The duty cycle is set by the ratio of the input voltage to
the output voltage. If the input voltage is 5V, and the output
voltage is 3.1V, the ratio of Vout/ Vin is 62%. Figure 13B
shows the result of a RC5050 going into its short circuit
mode with a duty cycle approximately of 20%. Calculating
the power in the MOSFET at each condition on the graph
(Figure 12) shows how the protection works. The power dis-
sipated in the MOSFET at normal operation for a load cur-
rent of 14.5A, is given by:
RC5050 and RC5051 Short Circuit Current
Characteristics
The RC5050 and RC5051 short circuit current characteristic
includes a hysteresis function that prevents the DC-DC con-
verter from oscillating in the event of a short circuit. Figure
12 shows the typical characteristic of the DC-DC converter
circuit with a 6mΩ sense resistor. The converter exhibits a
normal load regulation characteristic until the voltage across
the resistor exceeds the internal short circuit threshold of
120mV. At this point, the internal comparator trips and
signals the controller to turn off the gate drive to the power
MOSFET. This causes a drastic reduction in output voltage
as the load regulation collapses into the short circuit control
mode. The output voltage does not return to its nominal
value the output current is reduced to a value within the safe
range for the DC-DC converter.
2
14.5
˙
PD = I2 × RON × DutyCycle =
× .037 × .62 = 1.2W
---------
2
for each MOSFET.
3.5
The power dissipated in the MOSFET at short circuit
condition for a peak short current of 20A, is given by:
3.0
2.5
2.0
1.5
1.0
20
-----
PD
=
2 × .037 × .2 = 0.74W
2
for each MOSFET.
These calculations show that the MOSFET is not being
over-stressed during a short circuit condition.
0.5
0
0
5
10
15
20
25
Output Current
Figure 12. RC5050 Short Circuit Characteristic
13
AN50
APPLICATION NOTE
PD, Diode = IF, ave × VF × (1 – DutyCycle) =
14 × 0.45 × 0.8 ≈ 5W
Thus, for the Schottky diode, the thermal dissipation during
a short circuit is greatly magnified. This requires that the
thermal dissipation of the diode be properly managed by an
appropriate heat sink. To protect the Schottky from being
destroyed in the event of a short circuit, you should limit the
junction temperature to less than 130°C. You can find the
required thermal resistance using the equation for maximum
junction temperature:
T
J(max) – TA
PD = -------------------------------
RΘJA
Assuming that the ambient temperature is 50°C,
Figure 13A. V
Output Waveform for Normal
T
J(max) – TA
RΘJA = ------------------------------- = -------------------- = 16°C ⁄ W
PD
CCQP
130 – 50
Operation Condition with V = 3.3V@10A
out
5
Thus, you need to provide a heat sink that gives the Schottky
diode a thermal resistance of 16°C/W or lower to protect the
device during an indefinite short.
In summary, with proper heat sink, the Schottky diode is not
over-stressed during a short circuit condition.
Schottky Diode Selection
The application circuit diagram of Figure 3 shows a Schottky
diode, DS1. In non-synchronous mode, DS1 is used as a fly-
back diode to provide a constant current path for the inductor
when M1 is turned off. Table 10 shows the characteristics of
several Schottky diodes. Note that MBR2015CTL has a very
low forward voltage drop. This diode is ideal for applications
where the output voltage is required to be less than 2.8V.
Figure 13B. V
Output Waveform for
CCQP
Output Shorted to Ground
Table 10. Schottky Diode Selection Table
Power dissipation on the Schottky diode during a short cir-
cuit condition must also be considered. During normal oper-
ation, the Schottky diode dissipates power while the power
MOSFET is off. The power dissipated in the diode during
normal operation, is given by:
Manufacturer
Model #
Forward Voltage
VF
Conditions
Philips
PBYR1035
IF = 20A; Tj = 25°C
IF = 20A;Tj = 125°C
< 0.84v
< 0.72v
Motorola
MBR2035CT IF = 20A;Tj = 125°C
IF = 20A; Tj = 25°C
< 0.84v
< 0.72v
PD, Diode = IF × VF × (1 – DutyCycle) =
Motorola
MBR1545CT IF = 15A;Tj = 125°C
IF = 15A; Tj = 25°C
< 0.84v
< 0.72v
14.5 × 0.5V × (1 – 0.62) = 2.75W
Motorola
MBR2015CTL IF = 20A;Tj = 150°C
IF = 20A; Tj = 25°C
< 0.58v
< 0.48v
During a short circuit, the duty cycle dramatically reduces to
around 20%. The forward current in the short circuit condi-
tion decays exponentially through the inductor. The power
dissipated in the diode during short circuit condition, is
approximately given by:
Output Filter Capacitors
Output ripple performance and transient response are func-
tions of the filter capacitors. Since the 5V supply of a PC
motherboard may be located several inches away from the
DC-DC converter, the input capacitance may play an impor-
tant role in the load transient response of the RC5050 and
RC5051. The higher input capacitance, the more charge stor-
age is available for improving current transfer through the
1
1.5µs
– -------------
– -----------
IF, ending = Isc × e L ⁄ R = 20A × e 1.3µs ≈ 7.9A
IF, ave ≈ (20A + 7.9A) ⁄ 2 ≈ 14A
14
APPLICATION NOTE
AN50
FET. Low Equivalent Series Resistance (ESR) capacitors are
best suited for this type of application. Incorrect selection
can hinder the converter's overall performance. The input
capacitor should be placed as close to the drain of the FET as
possible to reduce the effect of ringing caused by long trace
lengths.
For I = 12.2A (0-13A load step) and ∆V = 100mV, the bulk
O
capacitance required can be approximated as follows:
IO × ∆T
12.2A × 2µs
C(µF) =-------------------------------------= ---------------------------------------------------------------= 2870µF
∆V – IO × ESR 100mV – 12.2A × 7.5mΩ
Because the control loop response of the controller is not
instantaneous, the initial load transient must be supplied
entirely by the output capacitors. The initial voltage deviation
is determined by the total ESR of the capacitors used and the
parasitic resistance of the output traces. For a detailed analysis
of capacitor requirements in a high-end microprocessor
system, please refer to Application Bulletin 5.
The ESR rating of a capacitor is a difficult number to
quantify. ESR is defined as the resonant impedance of the
capacitor. Since the capacitor is actually a complex imped-
ance device having resistance, inductance, and capacitance,
it is natural for this device to have a resonant frequency. As a
rule, the lower the ESR, the better suited the capacitor is for
use in switching power supply applications. Many capacitor
manufacturers do not supply ESR data. A useful estimate of
the ESR can be obtained using the following equation:
Input Filter
The DC-DC converter should include an input inductor
between the system +5V supply and the converter input as
described below. This inductor serves to isolate the +5V
supply from the noise in the switching portion of the
DC-DC converter, and to limit the inrush current into the
input capacitors during power up. A value of 2.5µH is rec-
ommended, as illustrated in Figure 14.
DF
ESR = ------------
2πfC
where DF is the dissipation factor of the capacitor, f is the
operating frequency, and C is the capacitance in farads.
With this in mind, correct calculation of the output capaci-
tance is crucial to the performance of the DC-DC converter.
The output capacitor determines the overall loop stability,
output voltage ripple, and load transient response. The calcu-
lation is as follows:
2.5µH
5V
Vin
1000µF, 10V
Electrolytic
0.1µF
IO × ∆T
C(µF) = -------------------------------------
∆V – IO × ESR
65-AP42-17
Figure 14. Input Filter
where ∆V is the maximum voltage deviation due to load
transients, ∆T is the reaction time of the power source (loop
response time for the RC5050 and RC5051 isapproximately
Bill of Material
2µs), and I is the output load current.
Table 11 is the Bill of Material for the Application Circuits
of Figure 3 and Figure 4.
O
Table 11. Bill of Materials for a 13A Pentium Pro Klamath Application
Quantity Reference
Manufacturer Part
Order #
Description
Requirements and
Comments
7
C4, C5, C7, Panasonic
C8, C9, C10, ECU-V1H104ZFX
C11
0.1µF 50V capacitor
1
1
1
3
4
1
1
C6
Panasonic
ECSH1CY475R
4.7µF 16V capacitor
120pF capacitor
Cext
Panasonic
ECU-V1H121JCG
C12
Panasonic
ECSH1CY105R
1µF 16V capacitor
C1, C2, C3
United Chemi-con
LXF16VB102M
1000µF 6.3V electrolytic
capacitor 10mm x 20mm
ESR < 0.047 Ω
C13, C14,
C15, C16
Sanyo
6MV1500GX
1500µF 6.3V electrolytic
capacitor 10mm x 20mm
ESR < 0.047 Ω
DS1
(note 1)
Motorola
MBR2015CT
Shottky diode, 15A
6.2V Zener Diode
Vf < 0.52V @ I = 10A
f
D1
Motorola 1N4691
15
AN50
APPLICATION NOTE
Table 11. Bill of Materials for a 13A Pentium Pro Klamath Application (continued)
Quantity Reference
Manufacturer Part
Order #
Description
Requirements and
Comments
1
1
L1
Pulse Engineering
PE-53680
1.3µH inductor
L2*
Pulse Engineering
PE-53681
2.5µH inductor
*Optional—helps
reduce ripple on 5v line
2-4
(note 2)
M1-M4
Rsense
R5
International Rectifier
IRF7413
N-Channel Logic Level
Enhancement Mode MOSFET
R
< 18mΩ
DS,ON
= 4.5V, I = 5A
V
GS
D
1
1
1
Coppel
CuNi Wire resistor
6 mΩ, 1W
Panasonic
ERJ-6GEY050Y
47Ω 5% resistors
10KΩ 5% resistor
R6
Panasonic
ERJ-6ENF10.0KY
U1
Raytheon
RC5050M or RC5051M
Programmable DC-DC
converter
Refer to Appendix A for Directory of component suppliers.
Notes:
1. When used in synchronous mode, a 1A schottky diode such as the 1N5817 should be substituted for the MBR2015CT.
2. A target R
value of 10mΩ should be used for each output driver switch. Refer to Table 3 for alternative MOSFETs.
DS,ON
trace and the large gate capacitance of the FET. This noise
PCB Layout Guidelines and
Considerations
radiates all throughout the board, and, because it is
switching at such a high voltage and frequency, it is very
difficult to suppress.
PCB Layout Guidelines
• Placement of the MOSFETs relative to the RC5050 is
critical. Place the MOSFETs (M1 & M2) so that the trace
length of the HIDRV pin from the RC5050 to the FET
gates is minimized. A long lead length on this pin would
cause high amounts of ringing due to the inductance of the
Figure 15 shows an example of good placement for the
MOSFETs in relation to the RC5050. In addition, this fig-
ure shows an example of problematic placement for the
MOSFETs.
M1
Good layout
Bad layout
M2
RC5050 10
RC5050 10
11
12
13
11
12
13
9
9
8
7
8
7
14
15
16
17
14
15
16
17
6
6
5
4
5
4
18
19
20
18
19
20
3
2
3
2
1
1
M1
M2
= “Quiet" Pins
Figure 15. Placement of the MOSFETs
16
APPLICATION NOTE
AN50
In general, all of the noisy switching lines should be kept
away from the quiet analog section of the RC5050. That is,
traces that connect to pins 12 and 13 (HIDRV and
VCCQP) should be kept far away from the traces that con-
nect to pins 1 through 5, and pin 16.
• Place the output bulk capacitors as close to the CPU as
possible to optimize their ability to supply instantaneous
current to the load in the event of a current transient.
Additional space between the output capacitors and the
CPU allows the parasitic resistance of the board traces to
degrade the DC-DC converter’s performance under severe
load transient conditions, causing higher voltage
• Place the 0.1µF decoupling capacitors as close to the
RC5050 pins as possible. Extra lead length negates their
ability to suppress noise.
deviation. For more detailed information regarding
capacitor placement, refer to Application Bulletin AB-5.
• Each VCC and GND pin should have its own via to the
appropriate plane. This helps to provide isolation between
pins.
• The traces that run from the RC5050 IFB (pin 4) and VFB
(pin 5) pins should be run next to each other and Kelvin
connected to the sense resistor. Running these lines
together prevents some of the common mode noise that is
presented to the RC5050 feedback input. Try, as much as
possible, to run the noisy switching signals (HIDRV &
VCCQP) on one layer, but use the inner layers for power
and ground only. If the top layer is being used to route all
of the noisy switching signals, use the bottom layer to
route the analog sensing signals VFB and IFB.
• Surround the CEXT timing capacitor with a ground trace.
Be sure to place a ground or power plane under the
capacitor for further noise isolation to provide additional
shielding to the oscillator pin 1 from the noise on the
PCB. In addition, place this capacitor as close to the
RC5050 pin 1 as possible.
• Place the MOSFETs, inductor and Schottky as close
together as possible for the same reasons on the first bullet
above. Place the input bulk capacitors as close to the
drains of MOSFETs as possible. In addition, placement of
a 0.1µF decoupling capacitor right on the drain of each
MOSFET helps to suppress some of the high frequency
switching noise on the input of the DC-DC converter.
Example of a PC Motherboard Layout and
Gerber File.
This section shows a reference design for motherboard
implementation of the RC5050 along with the Layout Gerber
File and Silk Screen. The actual PCAD Gerber File can be
obtained from Raytheon Electronics local Sales Office or
from the Semiconductor Division Marketing department at
415-966-7819.
17
AN50
APPLICATION NOTE
9. Next, look at HIDRV pin. This pin directly drives the
gate of the FET. It should provide a gate drive (Vgs) of
about 5V when turning the FET on. A careful study of
the layout is recommended. Refer to the “PCB Layout
Guidelines” section.
Guidelines for Debugging and
Performance Evaluations
DebuggingYour First Design Implementation
1. Note the setting of the VID pins to know what voltage is
to be expected.
10. Past experience shows that the most frequent errors are
incorrect components, improper connections, and poor
layout.
2. Do not connect any load to the circuit. While monitoring
the output voltage, apply power to the part with current
limiting at the power supply. This ensures that no cata-
strophic shorts are present.
Performance Evaluation
This section shows a sample evaluation results as a reference
guide for evaluating a DC-DC Converter using the RC5050
on a Pentium Pro motherboard.
3. If proper voltage is not achieved go to "Procedures "
below.
4. When you have proper voltage, increase the current lim-
iting of the power supply to 16A.
Load Regulation
5. Apply load at 1A increments. An active load (HP6060B
or equivalent) is suggested.
VID
Iload (A)
0.5
Vout (V)
3.0904
3.0825
3.0786
3.0730
3.0695
3.0693
3.0695
3.0695
3.0694
3.0694
3.0691
0.70%
10100
6. In case of poor regulation refer to "Procedures" below.
1.0
2.0
Procedures
3.0
1. If there is no voltage at the output and the circuit is not
drawing current look for openings in the connections,
check the circuitry versus schematic, and check the
power supply pins at the device to make sure that volt-
age(s) are applied.
4.0
5.0
6.0
7.0
2. If there is no voltage at the output and the circuit is
drawing excessive current (>100mA) with no load,
check for possible shorts. Determine the path of the
excessive current and which devise is drawing it—this
current may be drawn by peripheral components.
8.0
9.0
9.9
Load Regulation 0.5A – 9.9A
3. If the output voltage comes close to the expected value,
check the VID inputs at the device pins. The part is fac-
tory set to correspond to the VID inputs.
VID
Iload (A)
0.5
Vout (V)
3.2805
3.2741
3.2701
3.2642
3.2595
3.2597
3.2606
3.2611
3.2613
3.2611
3.2607
3.2599
3.2596
3.2596
0.64%
4. Premature shut down can be caused by an inappropriate
value of the sense resistor. See the “Sense Resistor” sec-
tion.
10010
1.0
2.0
5. Poor load regulation can be due to many causes. Check
the voltages and signals at the critical pins.
3.0
4.0
6. The VREF pin should be at the voltage set by the VID
pins. If the power supply pins and the VID pins are
correct the VREF should have the correct voltage.
5.0
6.0
7. Next check the oscillator pin.You should see a saw tooth
wave at the frequency set by the external capacitor.
7.0
8.0
8. When the VREF and CEXT pins are checked and
correct and the output voltage is incorrect, look at the
waveform at VCCQP. This pin should be swinging from
ground to +12V (in the +12V application), and from
slightly below +5V to about +10V (charge pump appli-
cation). If the VCCQP pin is noisy, with ripples/over-
shoots riding on it this may make the converter not to
function correctly.
9.0
10.0
11.0
12.0
12.4
Load Regulation 0.5A – 12.4A
18
APPLICATION NOTE
AN50
Output Voltage LoadTransients Due to Load Current Step
VID
Iload (A)
0.5
Vout (V)
2.505
2.504
2.501
2.496
2.493
2.493
2.492
2.492
2.491
2.490
2.989
2.488
2.486
2.485
2.484
0.84%
This test is performed using Intel P6.0/P6S/P6T Voltage
Transient Tester.
11010
1.0
Low to High 0.5A-9.9A
Current Step
-76.0mV
Refer to
2.0
Attachment
A for Scope
Picture
3.0
4.0
5.0
High to Low 9.9A-0.5A
Current Step
+70mV
Refer to
Attachment
B for Scope
Picture
6.0
7.0
8.0
Low to High 0.5A-12.4A -97.6mV
Current Step
Refer to
Attachment
C for Scope
Picture
9.0
10.0
11.0
12.0
13.0
13.9
High to Low 12.4A-0.5A +80.0mV
Current Step
Refer to
Attachment
D for Scope
Picture
Low to High 0.5A-13.9A -99.2mV
Current Step
Refer to
Load Regulation 0.5A – 13.9A
Attachment
E for Scope
Picture
Note:
Load regulation is expected to be typically around 0.8%. The
load regulation performance for this device under evaluation
is excellent.
High to Low 13.9A-0.5A +105.2mV Refer to
Current Step
Attachment
F for Scope
Picture
Note:
Transient voltage is recommended to be less than 4% of the
output voltage. The performance of the device under evalua-
tion is significantly better than a typical VRM.
19
AN50
APPLICATION NOTE
Input Ripple and Power on Input Rush Current
Power on Input Rush Current was not measured on the moth-
erboard because we did not want to cut the 5V trace and
insert a current probe in series with the supply. However,
with the input filter design, the Input Rush Current is well
within specification.
Iload = 9.9A Input Ripple
Voltage = 15mV ment G for Scope
Picture
Refer to Attach-
Note:
Excellent input ripple voltage. Input ripple voltage is recom-
mended to be less than 5% of the output voltage.
Component Case Temperature
Case Temperature
Iload = 9.9A
Case Temperature Case Temperature
Iload = 12.4A
Iload = 13.9A
Device
Description
(°C)
(°C)
(°C)
Q3A
MOSFET
K1388
57
63
64
56
70
66.3
66.6
61.2
87
Q3B
L1
MOSFET
K1388
58
53
66
Inductor,
Unknown
Q2
Schottky Diode
2048CT
IC
Raytheon’s RC5050
52
38.2
35
54
58
39
Cin
Cout
Input Cap. 1000µF
36.8
34.8
Output Cap.
38.2
1500µF
Note:
The values for case temperatures are within guidelines. That is, case temperatures for all components should be below
105°C @25°C Ambient.
Evaluation Summary
The on-board DC-DC converter is fully functional. It has
excellent load regulation, transient response, and input volt-
age ripple.
Attachment B
Attachment A
20
APPLICATION NOTE
AN50
Attachment C
Attachment E
Attachment F
Attachment D
Attachment G
21
AN50
APPLICATION NOTE
Summary
RC5050 Evaluation Board
This application note covers many aspects of the RC5050
and RC5051 for implementation of a DC-DC converter a on
Pentium Pro motherboard. A detailed discussion includes
the processor power requirements, a description of the
RC5050 and RC5051, design considerations and compo-
nents selection, layout guidelines and considerations, guide-
lines for debugging, and performance evaluations.
Raytheon Electronics provides an evaluation board to verify-
ing system level performance of the RC5050. The evaluation
board serves as a guide to performance expectations when
using the supplied external components and PCB layout.
Call Raytheon Electronics local Sales Office or the Market-
ing department at 415-966-7819 for an evaluation board.
22
APPLICATION NOTE
AN50
Appendix A
Directory of Component Suppliers
Dale Electronics, Inc.
E. Hwy. 50, PO Box 180
Yankton, SD 57078-0180
PH: (605) 665-9301
National Semiconductor
2900 Semiconductor Drive
Santa Clara, CA 95052-8090
PH: (800) 272-9959
Fuji Electric
Collmer Semiconductor Inc.
14368 Proton Rd.
Dallas, Texas 75244
PH: (214)233-1589
Nihon Inter Electronics Corp.
Quantum Marketing Int’l, Inc.
12900 Rolling Oaks Rd.
Caliente, CA 93518
PH: (805) 867-2555
General Instrument
Panasonic Industrial Co.
6550 Katella Avenue
Cypress, CA 90630
PH: (714) 373-7366
Power Semiconductor Division
10 Melville Park Road
Melville, NY 11747
PH: (516) 847-3000
Pulse Engineering
Hoskins Manufacturing Co.
(Copel Resistor Wire)
10776 Hall Road
12220 World Trade Drive
San Diego, CA 92128
PH: (619) 674-8100
Hamburg, MI 48139-0218
PH: (313) 231-1900
Sanyo Energy USA
2001 Sanyo Avenue
San Diego, CA 92173
PH: (619) 661-6620
Intel Corp.
5200 NE Elam Young Pkwy.
Hillsboro, OR. 97123
PH: (800) 843-4481 Tech. Support
for Power Validator
Siliconix
Temic Semiconductors
2201 Laurelwood Road
Santa Clara, CA 95056-1595
PH: (800) 554-5565
International Rectifier
233 Kansas St.
El Segundo, CA 90245
PH: (310) 322-3331
Sumida Electric USA
5999 New Wilke Road Suite #110
Rolling Meadows, IL 60008
PH: (708) 956-0702
IRC Inc.
PO Box 1860
Boone, NC 28607
PH: (704) 264-8861
Xicon Capacitors
PO Box 170537
Motorola Semiconductors
PO Box 20912
Arlington, Texas 76003
PH:(800) 628-0544
Phoenix, Arizona 85036
PH:(602) 897-5056
23
AN50
APPLICATION NOTE
LIFE SUPPORT POLICY
FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES
OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR
CORPORATION. As used herein:
1. Life support devices or systems are devices or systems
which, (a) are intended for surgical implant into the body,
or (b) support or sustain life, and (c) whose failure to
perform when properly used in accordance with
instructions for use provided in the labeling, can be
reasonable expected to result in a significant injury of the
user.
2. A critical component in any component of a life support
device or system whose failure to perform can be
reasonably expected to cause the failure of the life support
device or system, or to affect its safety or effectiveness.
Fairchild Semiconductor
Corporation
Fairchild Semiconductor
Europe
Fairchild Semiconductor
Hong Kong Ltd.
National Semiconductor
Japan Ltd.
Americas
Fax: +49 (0) 1 80-530 85 86
13th Floor, Straight Block,
Ocean Center, 5 Canto Rd.
Tsimshatsui, Kowloon
Hong Kong
Tel:81-3-5620-6175
Fax:81-3-5620-6179
Customer Response Center
Tel:1-888-522-5372
Email: [email protected]
Deutsch Tel: +49 (0) 8 141-35-0
English Tel: +44 (0) 1 793-85-68-56
Italy
Tel: +39 (0) 2 57 5631
Tel:+852 2737-7200
Fax:+852 2314-0061
2/98 0.0m
Stock#AN30000050
1998 Fairchild Semiconductor Corporation
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